Vector controller for permanent-magnet synchronous electric motor

ABSTRACT

A method of controlling a current command by comparing voltage with a set value needs to vary the set value depending on voltage fluctuation, which involves taking a complicated control. A vector controller for a permanent-magnet synchronous electric motor, according to the present invention, can realize with a simplified configuration a field-weakening operation in a one-pulse mode in a high speed range by providing a current command compensator that corrects a current command by a corrected current command calculated based on a modulation index.

TECHNICAL FIELD

The present invention relates to vector controllers for permanent-magnetsynchronous electric motors.

BACKGROUND ART

Vector control technologies for permanent-magnet synchronous electricmotors (hereinafter abbreviated as “electric motor”) using an inverterare widely employed in industry. By separately controlling the magnitudeand the phase of inverter output voltage, a current vector in anelectric motor is optimally controlled, so that torque of the electricmotor is fast and instantaneously controlled. Permanent-magnetsynchronous electric motors are known as high-efficiency electric motorsin comparison with induction motors because no energizing current isneeded due to establishment of magnetic field by the permanent magnetand no secondary copper loss is generated due to no rotor current. Forthat reason, application of permanent-magnet synchronous electric motorsto electric railcars has been investigated in recent years.

Subjects with controllers in applying permanent-magnet synchronouselectric motors to electric railcars are to realize a stablefield-weakening operation up to a high speed range and to achieve astable transition to a one-pulse mode in which inverter loss can beminimized and voltage applied to the electric motors can be maximized.The one-pulse mode is an operation mode for inverters, in which aninverter outputs, as its output line voltage, square waves having apositive and a negative maximum rectangular voltages of 120 degreedurations each that are repeated one after another with a zero voltageperiod of 60 degrees therebetween, in one cycle, i.e., 360 degrees.

The following method is disclosed in Patent Document 1 as a relatedprior art. A voltage setting unit is provided that receives a voltagefixing command and a voltage command calculated based on a currentcommand. When the voltage fixing command is input, the voltage settingunit outputs a voltage command as a new voltage command by setting itsmagnitude to a predetermined voltage set value. Amagnetic-field-direction (d-axis) current command is then correctedusing a magnetic-field-direction (d-axis) current correcting valueobtained by taking a proportional-integral control of the differencebetween the voltage command calculated from the current command and thenew voltage command. A modulation index for the inverter is thencalculated from the voltage command to control the inverter, so that afield-weakening operation is performed.

Patent Document 1: Japan Patent Application Laid-Open No. H09-84399 (seeparas. [0023]-[0029]).

DISCLOSURE OF THE INVENTION

In Patent Document 1 cited above, however, how to generate the voltagefixing command is not disclosed and the voltage setting unit needs to beprovided anew. Moreover, a capacitor voltage always fluctuates, so thata maximum voltage that the inverter can output also fluctuatesaccordingly. In order to maximize voltage applied to the electric motoraccording to the method disclosed in Patent Document 1, it is necessaryto vary a timing of generating the voltage fixing command and to vary avoltage set value, depending on the fluctuation in the capacitorvoltage, which involves taking a complicated control.

Furthermore, a value is used as the field-direction (d-axis) currentcorrecting value that is obtained by taking the proportional-integralcontrol of the deviation between the voltage command calculated based ona current command and the new voltage command whose magnitude is set bythe voltage fixing command. Accordingly, when the deviation between thevoltage command and the new voltage command is not zero, i.e., during aninput to the proportional-integral control remains not zero, the controloperation continuously accumulates an integration value. For thatreason, when the voltage command theoretically calculated based on thecurrent command cannot be set to a value smaller than the voltagecommand set anew—for example, when a torque command is excessively largefor rotation speed of the electric motor—even though amagnetic-field-direction current is corrected using themagnetic-field-direction current correcting value, the differencebetween the voltage command and the voltage command set anew cannot beset to zero and the integration value in the proportional-integralcontrol is continuously accumulated, so that themagnetic-field-direction current correcting value excessively increasesas time elapses. When the magnetic-field-direction current correctingvalue becomes excessively large, the vector control cannot be normallyperformed. A complicated operation is therefore required in a practiceuse, such as limiting the integration value to a value less than anupper limit or resetting the integration value under a specifiedcondition.

The present invention is made to solve the above-described problems, andprovides a vector controller for a permanent-magnet synchronous electricmotor that can realize with a simplified configuration a stableone-pulse-mode field-weakening control in a high speed range.

Means for Solving the Problem

A vector controller for a permanent-magnet synchronous electric motor,according to the present invention, controls an alternating current froman inverter that drives the permanent-magnet synchronous electric motorso as to come into coincidence with a current command, and provided witha reference phase-angle calculation unit for generating a referencephase angle of the permanent-magnet synchronous electric motor; acurrent command generation unit for generating the current command usinga given torque command; a current control unit for making a controlcalculation of a current error between the current command and a currentthrough the permanent-magnet synchronous electric motor, to output thecalculated current error; a decoupling voltage calculation unit forcalculating a feed-forward voltage using motor parameters of thepermanent-magnet synchronous electric motor and the current command; amodulation index calculation unit for outputting a modulation index forthe inverter by receiving a direct-current voltage to the inverter and avoltage command that is the sum of the current error and thefeed-forward voltage; a control phase-angle calculation unit foroutputting a control phase angle for the inverter by receiving thevoltage command and the reference phase angle; a pulse-width-modulationsignal generation unit for generating pulse-width-modulation signals forthe inverter using the modulation index and the control phase angle; anda current command compensator for correcting the current command using acorrected current command calculated based on the modulation index;wherein the current command compensator sets the corrected currentcommand to a value obtained by processing through a time delay elementand by multiplying by a predetermined constant the difference betweenthe modulation index and a predetermined modulation index set value.

Effects of the Invention

A vector controller for a permanent-magnet synchronous electric motor,according to the invention, controls an alternating current from aninverter that drives the permanent-magnet synchronous electric motor soas to come into coincidence with a current command, and provided with areference phase-angle calculation unit for generating a reference phaseangle of the permanent-magnet synchronous electric motor; a currentcommand generation unit for generating the current command using a giventorque command; a current control unit for making a control calculationof a current error between the current command and a current through thepermanent-magnet synchronous electric motor, to output the calculatedcurrent error; a decoupling voltage calculation unit for calculating afeed-forward voltage using motor parameters of the permanent-magnetsynchronous electric motor and the current command; a modulation indexcalculation unit for outputting a modulation index for the inverter byreceiving a direct-current voltage to the inverter and a voltage commandthat is the sum of the current error and the feed-forward voltage; acontrol phase-angle calculation unit for outputting a control phaseangle for the inverter by receiving the voltage command and thereference phase angle; a pulse-width-modulation signal generation unitfor generating pulse-width-modulation signals for the inverter using themodulation index and the control phase angle; and a current commandcompensator for correcting the current command using a corrected currentcommand calculated based on the modulation index; wherein the currentcommand compensator sets the corrected current command to a valueobtained by processing through a time delay element and by multiplyingby a predetermined constant the difference between the modulation indexand a predetermined modulation index set value. Therefore, an effect isbrought about that can realize with a simplified configuration a stableone-pulse-mode field-weakening control in a high speed range.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating an example of a configuration ofa vector controller for a permanent-magnet synchronous electric motor,according to Embodiment 1 of the present invention;

FIG. 2 is a block diagram illustrating an example of a configuration ofa current command generation unit in Embodiment 1 of the invention;

FIG. 3 is a block diagram illustrating an example of a configuration ofa PWM-signal generation unit in Embodiment 1 of the invention;

FIG. 4 shows charts for illustrating a modulation index PMF, pulse-modetransitions, switching operations, and a control-mode transition, withinverter angular frequency ω, in Embodiment 1 of the invention;

FIG. 5 is a block diagram illustrating an example of a configuration ofa current command compensator in Embodiment 1 of the invention;

FIG. 6 is a graph showing a relation of the deviation between the sum ofsquares of dq-axis current commands and that of squares of dq-axiscurrents, to a d-axis current error, in Embodiment 1 of the invention;

FIG. 7 is a graph showing a relation of the deviation between themagnitude of current command vectors and that of current vectors, to thed-axis current error, in Embodiment 1 of the invention;

FIG. 8 is a graph showing a relation of the deviation between the sum ofsquares of dq-axis current commands, and that of squares of dq-axiscurrent commands, to a q-axis current error, in Embodiment 1 of theinvention;

FIG. 9 is a graph showing a relation of the deviation between themagnitude of current command vectors and that of current vectors, to theq-axis current error, in Embodiment 1 of the invention;

FIG. 10 illustrates charts showing simulated operating waveforms oftorque commands, torques, d-axis current commands, d-axis currents,q-axis current commands, and q-axis currents, in Embodiment 1 of theinvention; and

FIG. 11 illustrates charts showing simulated operating waveforms ofmodulation indexs, corrected current commands, U-phase voltage commands,synchronous three-pulse PWM mode flags, synchronous one-pulse modeflags, and U-phase currents, in Embodiment 1 of the invention.

REFERENCE NUMERALS

-   1: capacitor,-   2: inverter,-   3, 4, 5: current sensor,-   6: electric motor,-   7: resolver,-   8: voltage sensor,-   10: current command generation unit,-   11: d-axis fundamental current command generation unit,-   14: adder,-   15: q-axis current command generation unit,-   20: d-axis current control unit,-   21: q-axis decoupling calculation unit, (decoupling calculation    unit),-   22: d-axis decoupling calculation unit (decoupling calculation    unit),-   23: q-axis current control unit,-   30: modulation index calculation unit,-   40: control phase-angle calculation unit,-   50: PWM signal generation unit,-   53: multiplier,-   54: gain adjustment table,-   55: voltage command calculation unit,-   57: multi-pulse carrier-signal generation unit,-   58: synchronous three-pulse carrier-signal generation unit,-   59: switch,-   60: pulse-mode switching process unit,-   61, 62, 63: comparator,-   64, 65, 66: NOT-circuit,-   70: inverter angular-frequency calculation unit,-   80: current command compensator,-   81: limiter,-   82: first-order delay element,-   83: proportional gain element,-   85: parameter error correction unit,-   90: three-phase to dq-axis coordinate transform unit,-   95: reference phase-angle calculation unit, and-   100: vector controller.

BEST MODE FOR CARRYING OUT THE INVENTION Embodiment 1

FIG. 1 is a block diagram illustrating an example of a configuration ofa vector controller for a permanent-magnet synchronous motor, accordingto Embodiment 1 of the present invention. As shown in FIG. 1, a maincircuit is configured with a capacitor 1 that is a direct-current powersource, an inverter 2 that converts direct-current voltage of thecapacitor 1 into an alternating-current voltage of any given frequency,and a permanent-magnet synchronous electric motor 6 (hereinafter, simplyreferred to as “electric motor”).

The main circuit is also provided with a voltage sensor 8 that sensesvoltage of the capacitor 1, current sensors 3, 4, and 5 that sensecurrents iu, iv, and iw through the output lines of the inverter 2. Theelectric motor 6 is provided with a resolver 7 that senses a rotormechanical angle θm. Each of these sensing signals is input into avector controller 100.

The resolver 7 may be substituted with an encoder, or a positionsensorless method may be used in which a position signal is calculatedfrom a sensed voltage, current, or the like instead of a position signalobtained by the resolver 7. In these cases, the resolver 7 isunnecessary, in other words, acquisition of a position signal is notlimited to using the resolver 7.

As for the current sensors 3, 4, and 5, a configuration may be employedin which sensors are provided for at least two phase-lines since acurrent through the other phase line can be determined by calculation,or respective currents are determined by simulating output currents ofthe inverter 2 from a current on the direct-current side thereof.

The inverter 2 receives gate signals U, V, W, and X, Y, Z that aregenerated by the vector controller 100 to take a pulse-width-modulation(PWM) control of switching elements built in the inverter 2. A PWMvoltage-source inverter is suitable for the inverter 2, and since itsconfiguration is publicly known, a detailed description thereof isomitted.

The vector controller 100 receives a torque command T* from a externalcontroller, not shown, and controls the inverter 2 so that torque Tproduced by the electric motor 6 comes into coincident with the torquecommand T*.

Next, a configuration of the vector controller 100 is described. Thevector controller 100 is configured with a reference phase-anglecalculation unit 95 that calculates a reference phase angle θe from therotor mechanical angle θm; a three-phase to dq-axis coordinate transformunit 90 that generates a d-axis current id and a q-axis current iq fromthe three phase currents iu, iv, and iw sensed by the current sensors 3,4, and 5, respectively, and from the reference phase angle θe; aninverter angular-frequency calculation unit 70 that calculates aninverter angular frequency ω from the reference phase angle θe; acurrent command generation unit 10 that generates a d-axis currentcommand id* and a q-axis current command iq* from the torque command T*input from externally and a later-described corrected current commanddV*; a d-axis current control unit 20 that generates a d-axis currenterror pde by taking a proportional-integral control of the differencebetween the d-axis current command id* and the d-axis current id; aq-axis current control unit 23 that generates a q-axis current error pqeby taking a proportional-integral control of the difference between theq-axis current command iq* and the q-axis current iq; a q-axisdecoupling calculation unit 21 that calculates a q-axis feed-forwardvoltage VqFF from the d-axis current command id* and the inverterangular frequency ω; a d-axis decoupling calculation unit 22 thatcalculates a d-axis feed-forward voltage VdFF from the q-axis currentcommand iq* and the inverter angular frequency ω; a modulation indexcalculation unit 30 that calculates a modulation index PMF, a controlphase-angle calculation unit 40 that calculates a control phase angle θfrom a d-axis voltage command Vd* that is the sum of the d-axis currenterror pde and the d-axis feed-forward voltage VdFF, a q-axis voltagecommand Vq* that is the sum of the q-axis current error pqe and theq-axis feed-forward voltage VqFF, the reference phase angle θe, and froma later-described control phase-angle correcting value dTHV; aPWM-signal generation unit 50 that generates the gate signals U, V, W,and X, Y, Z for the inverter 2; a current command compensator 80 forcalculating the corrected current command dV by receiving the modulationindex PMF; and a parameter-error correction unit 85 that calculates thecontrol phase-angle correcting value dTHV from the d-axis current id,the q-axis current iq, the d-axis current command id*, and the q-axiscurrent command iq*.

Here, the modulation index calculation unit 30 receives the d-axisvoltage command Vd* that is the sum of the d-axis current error pde andthe d-axis feed-forward voltage VdFF, the q-axis voltage command Vq*that is the sum of the q-axis current error pqe and the q-axisfeed-forward voltage VqFF, the reference phase angle θe, and a voltageEFC of the capacitor 1. The PWM-signal generation unit 50 receives themodulation index PMF and the control phase angle θ.

Next, detailed configurations of each of the control blocks mentionedabove will be described. The reference phase-angle calculation unit 95calculates from the rotor mechanical angle θm the reference phase-angleθe that is an electric angle, based on the following equation (1):θe=θm*PP  (1),where PP denotes a pole pair number of the electric motor 6.

The three-phase to dq-axis coordinate transform unit 90 generates thed-axis current id and q-axis current iq from the three phase currentsiu, iv, and iw and the reference phase-angle θe, based on the followingequation (2):

$\begin{matrix}{\begin{pmatrix}{iq} \\{id}\end{pmatrix} = {\sqrt{\frac{2}{3}}{\begin{pmatrix}{{\cos\;\theta\; e\;{\cos\left( {{\theta\; e} - {\frac{2}{3}\pi}} \right)}{\cos\left( {{\theta\; e} + {\frac{2}{3}\pi}} \right)}}\;} \\{\sin\;\theta\; e\;{\sin\left( {{\theta\; e} - {\frac{2}{3}\pi}} \right)}{\sin\left( {{\theta\; e} + {\frac{2}{3}\pi}} \right)}}\end{pmatrix} \cdot \begin{pmatrix}{iu} \\{iv} \\{iw}\end{pmatrix}}}} & (2)\end{matrix}$

The inverter angular-frequency calculation unit 70 calculates theinverter angular frequency ω by differentiating the reference phaseangle θe, based on the following equation (3):ω=dθe/dt  (3).

A configuration of the current command generation unit 10 is described.FIG. 2 is a block diagram illustrating an example of a configuration ofthe current command generation unit 10 in Embodiment 1 of the invention.A d-axis fundamental current command generation unit 11 receives thetorque command T*, to generate a d-axis fundamental current commandid1*. A maximum torque control method that can generate a desired torqueof the electric motor 6 with a minimum current is known as a method ofgenerating the d-axis fundamental current command id1*, in which anoptimum d-axis fundamental current command id1* is obtained byreferencing a map on the basis of the torque command T* or by using anarithmetic expression. Since the unit can be configured by using knownart, a detailed description of the unit is omitted here.

After the d-axis fundamental current command id1* is generated, thed-axis current command id* is then obtained by adding the correctedcurrent command dV to the d-axis fundamental current command id1* by anadder 14. The corrected current command dV is provided for so-calledfield-weakening control. The correcting value dV has a negative value tocorrect the value id1* in the negative direction, so that the d-axiscurrent command id* increases in the negative direction; wherebymagnetic field is generated in such a direction as to cancel magneticfield produced by the permanent magnet of the electric motor 6, so thatflux linkage of the electric motor 6 is weakened, that is, afield-weakening control is realized. Since a method of generating thecorrected current command dV is a key feature of the invention, it willbe described later.

The q-axis current command iq* is finally generated by a q-axis currentcommand generation unit 15 from the d-axis current command id* and thecommand torque vale T*. As for a method of generating the q-axis currentcommand, there has also been a method in which an optimum q-axis currentcommand iq* is obtained by referencing a map or by using an arithmeticexpression, as mentioned above. Since the generation unit can beconfigured by using known art, a detailed description of the unit isomitted here.

The q-axis current control unit 23 and the d-axis current control unit20 generate the q-axis current error pqe and the d-axis current errorpde obtained by proportional-integral amplification of the differencesbetween the q-axis current command iq* and the q-axis current, andbetween the d-axis current command id* and the d-axis current, based onthe following equations (4) and (5), respectively:pqe=(K ₁ +K ₂ /s)*(iq*−iq)  (4) andpde=(K ₃ +K ₄ /s)*(id*−id)  (5),where K₁ and K₃ are proportional gains, and K₂ and K₄ are integralgains.

As described later, the q-axis current error pqe and the d-axis currenterror pde are gradually decreased to zero after a transition from acontrol mode 1 (described later) to a control mode 2 (described later),and are gradually increased when transition is made from the controlmode 2 to the control mode 1 on the contrary.

The d-axis decoupling calculation unit 22 and the q-axis decouplingcalculation unit 21 that are decoupling voltage calculation unitscalculate the d-axis feed-forward voltage VdFF and the q-axisfeed-forward voltage VqFF, based on the following equations (6) and (7),respectively:VdFF=(R ₁ +s*Ld)*id*−ω*Lq*iq*   (6) andVqFF=(R ₁ +s*Lq)*iq*+ω*(Lq*id*+φa)  (7),where R₁ denotes primary winding resistance (Ω), Ld and Lq denote d-axisinductance (H) and q-axis inductance (H), respectively, φa, magneticflux of the permanent magnet (Wb), and s, the differential operator.

The modulation index PMF here denotes a ratio of command output-voltagevector magnitude VM* for the inverter to the maximum voltage VM_(max)that the inverter is able to output, and PMF=1.0 indicates that thecommand output-voltage vector magnitude VM* is equal to the maximumvoltage VM_(max).

By thus defining the modulation index PMF, the modulation index PMFbecomes zero when the command output-voltage vector magnitude VM* iszero and becomes 1.0 under the condition that the inverter outputs itsmaximum voltage. An output ratio of inverter voltage is thereby easy tonotice intuitively, which has an advantage that makes it easy toconstruct and set control processes in which the modulation index PMF isreferred to, such as later-described switching processes of pulse modesand control modes.

The modulation index calculation unit 30 calculates, according to thebefore-mentioned definition of the modulation index PMF, the modulationindex PMF from the d-axis voltage command Vd* that is the sum of thed-axis current error pde and the d-axis feed-forward voltage VdFF, theq-axis voltage command Vq* that is the sum of the q-axis current errorpqe and the q-axis feed-forward voltage VqFF, the reference phase angleθe, and the voltage EFC of the capacitor 1, based on the followingequation (8):PMF=VM*/VM _(max)   (8),whereVM _(max=)(SQRT(6)/π)*EFC   (9) andVM*=SQRT(Vd* ² +Vq* ²)  (10).

The control phase-angle calculation unit 40 calculates the control phaseangle θ from the d-axis voltage command Vd* that is the sum of thed-axis current error pde and the d-axis feed-forward voltage VdFF, theq-axis voltage command Vq* that is the sum of the q-axis current errorpqe and the q-axis feed-forward voltage VqFF, the reference phase angleθe, and the control phase-angle correcting value dTHV, based on thefollowing equation (11):θ=θe+π+THV+dTHV   (11),whereTHV=tan⁻¹(Vd*/Vq*)  (12).

Next, a configuration of the PWM-signal generation unit 50 is described.FIG. 3 is a block diagram illustrating an example of a configuration ofthe PWM-signal generation unit 50 in Embodiment 1 of the invention. Asshown in FIG. 3, a command-voltage generation unit 55 generates from themodulation index PMF and the control phase-angle θ a command U-phasevoltage value Vu*, a command V-phase voltage value Vv*, and a commandW-phase voltage value VW* that are command three phase voltages, basedon the following equations (13) through (15):Vu*=PMFM* sin θ  (13),Vv*=PMFM* sin (θ−(2*π/3)  (14), andVw*=PMFM* sin (θ−(4*π/3)  (15).

Here, the coefficient PMFM is command voltage amplitude obtained bymultiplying a modulation index PMF by an output of a gain adjustmenttable 54 by a multiplier 53. The gain adjustment table 54 is forcorrecting the difference between a relation of an inverter outputvoltage VM to the modulation index PMF in a multi-pulse PWM mode andthat in a synchronous three-pulse PWM mode. The correction is outlinedas follows.

The maximum voltage (RMS value) that the inverter can output withoutdistortion is 0.612*EFC in the multi-pulse PWM mode and isVM_(max)(=0.7797*EFC) in the synchronous three-pulse PWM mode. Namely,an output voltage of the inverter for a modulation index PMF in themulti-pulse PWM mode becomes 1/1.274, compared with that in thesynchronous three-pulse PWM mode. In order to compensate the difference,a modulation index PMF in the multi-pulse PWM mode is multiplied by1.274 to be output as the command voltage amplitude PMFM into theabove-mentioned command-voltage calculation unit 55.

The command U-phase voltage value Vu*, the command V-phase voltage valueVv*, and the command W-phase voltage value Vw* are then compared inmagnitude with carrier signals CAR by comparators 61, 62, and 63, sothat the gate signals U, V, and Ware generated, respectively. The gatesignals X, Y, and Z are also generated from the gate signals U, V, and Wvia NOT-circuits 64, 65, and 66, respectively. The carrier signals CARare signals each selected, via a switch 59 by a pulse-mode switchingprocess unit 60, among a multi-pulse carrier signal A (around 1 kHz ingeneral) generated in a multi-pulse carrier-signal generation unit 57, asynchronous three-pulse carrier signal B generated in a synchronousthree-pulse carrier-signal generation unit 58, and a zero value Cemployed in a one-pulse mode. The multi-pulse carrier signal A and thesynchronous three-pulse carrier signal B each take values ranging from−1.0 to 1.0 centered at zero.

In addition, the pulse-mode switching process unit 60, depending on amodulation index PMF and a control phase angle θ, operates to change theswitch 59 to the asynchronous carrier A side when the modulation indexPMF is in a low range (equal to or lower than 0.785), to the synchronousthree-pulse carrier B side when it is in a range from 0.785 to lowerthan 1.0, and to the zero value C side, when it reaches 1.0,respectively.

Such configuration allows a pulse mode to be automatically changed tothe one-pulse mode at the timing when a modulation index PMF equals 1.0,and, on the contrary, to be automatically changed to the synchronousthree-pulse mode when a modulation index PMF becomes lower than 1.0.Namely, it is possible to easily vary output voltage of the inverter 2from the minimum to the maximum.

While the threshold for a modulation index PMF is set to the value of0.785 at which the asynchronous carrier and the synchronous three-pulsecarrier are switched, the threshold may be smaller than this value.

The synchronous three-pulse PWM mode here referred to is a pulse modenecessary for outputting a voltage corresponding to a modulation indexPMF equal to or higher than 0.785 which voltage is impossible to beoutput in the multi-pulse PWM mode. Although such a voltagecorresponding to the synchronous three-pulse mode can also be output byemploying a configuration such that an over-modulation method is used ina multi-pulse PWM mode, a synchronous five-pulse mode, a synchronousnine-pulse mode, or the like, an output of the inverter 2 becomessignificantly nonlinear with a modulation index PMF, which risesnecessity for correcting the nonlinearity, making the configurationcomplicated.

Each of the calculation equations shown above is generally processed bya microcomputer using software. However, when the calculation is madewith low precision (small bit count) to reduce calculating loads on themicrocomputer, a modulation index PMF does not exactly reach 1.0 but mayhave a value smaller than it such as 0.999 at the timing when commandoutput-voltage vector magnitude VM* for the inverter becomes the maximumvoltage VM_(max). In this case, although some voltage jump is involved,it is practicable that a pulse mode is changed to the one-pulse modewhen a modulation index PMF becomes, for example, 0.95 or larger.

Furthermore, a timing of changing a pulse mode may be finely adjusted byusing the control phase angle θ. By the fine adjustment, current throughthe electric motor can be prevented from rippling at changing the pulsemode.

FIG. 4 shows charts for illustrating the modulation rate PMF,transitions over the pulse modes, operations of the switch 59, and atransition over the control modes, with the inverter angular frequencyω, in Embodiment of the invention. As shown in FIG. 4, when electricrailcars run at a low speed, i.e., the inverter angular frequency ω islow, a modulation index PMF is small and the switch 59 is selected tothe A side, i.e., a pulse mode is set to the multi-pulse PWM mode. Atthe same time, a control mode is set to the control mode 1, and theq-axis current control unit 23 and the d-axis current control unit 20operate according to the above equations (4) and (5), respectively. Whenthe modulation index PMF equals to or exceeds 0.785 with increasingelectric railcar speed, since an output voltage of the inverter 2 issaturated in the multi-pulse PWM mode, the switch 59 is changed to the Bside to set the pulse mode to the synchronous three-pulse PWM mode.

At the same time, the control mode is selected to the control mode 2,and the d-axis current control unit 20 and the q-axis current controlunit 23 cease their calculations so that their outputs are reduced tozero. The reason for reducing to zero is as follows. In the synchronousthree-pulse PWM mode, the number of pulses per half cycle of outputvoltage of the inverter decreases from ten or more in the multi-pulsePWM mode to three, so that a control delay increases. If the d-axiscurrent control unit 20 and the q-axis current control unit 23 are leftto continue the calculations, there is a fear that the control systembecomes unstable. Therefore, the d-axis current control unit 20 and theq-axis current control unit 23 are stopped to make their calculations.

In addition, it is preferable for avoiding a shock produced at the modechange that the outputs of the d-axis current control unit 20 and theq-axis current control unit 23 be gradually reduced toward zero with apredetermined time constant during the reduction process.

In control mode 2, a mismatch between electric motor parameters andcontrol parameters arises from stopping the calculations of the d-axiscurrent control unit 20 and the q-axis current control unit 23.Deviations of a torque of and a current through the electric motor 6from their command values are generated from the mismatch and like.Control errors such as the deviations can be suppressed by correctingthe control phase angle θ using the control phase-angle correcting valuedTHV generated by the parameter-error correction unit 85 using thed-axis current id, the q-axis current iq, the d-axis current commandid*, and the q-axis current command iq*. A detailed configuration of theparameter-error correction unit 85 will be described later.

In addition, the output of the parameter error correction unit 85 isincreased after the change from the control mode 1 to the control mode2, and decreased to zero after the change from the control mode 2 to thecontrol mode 1 on the contrary. The increase and the decrease arepreferably performed slowly with a predetermined time constant. Anunstable control can thereby be avoided that results from competition ofthe output of the d-axis current control unit 20 or the q-axis currentcontrol unit 23 with that of the parameter-error correction unit 85.

When the modulation index PMF becomes 1.0 or larger with electricrailcar speed being further increased, the pulse mode is changed to theone-pulse mode by changing the switch 59 to the C side. The control modestill remains the control mode 2. When the electric railcars slow downby regenerative brakes, which is not shown, the pulse mode is transitedfrom the one-pulse mode to the multi-pulse PWM mode through thesynchronous three-pulse PWM mode and the switch 59 is changed from the Cside to the A side through the B side in a sequence reverse to theabove, so that the control mode is transited from the control mode 2 tothe control mode 1.

Next, a description is made of a configuration of the current commandcompensator 80 that is a key component to demonstrate effects of theinvention. FIG. 5 is a block diagram illustrating an example of aconfiguration of the current command compensator 80 in Embodiment 1 ofthe invention. As shown in FIG. 5, the difference between a modulationindex set value PMF_(max) and a modulation index PMF is input into alimiter 81 that is able to limit the difference to the range between anupper limit and a lower limit. The limiter 81 is configured so as to beable to limit its input signal to the range between an upperdeviation-limit set value LIMH and a lower deviation-limit set valueLIML, to output the limited signal. The output of the limiter 81 isinput into a first-order delay element 82. An output of the first-orderdelay element 82 is input into a proportional gain element 83 andmultiplied by a gain K that is a predetermined coefficient, to be outputas the corrected current command dV. With the first-order delay element82, even when the difference between the modulation index set valuePMF_(max) and the modulation index PMF upsurges, the corrected currentcommand dV increases with a predetermined time constant.

As described above, the corrected current command dV is expressed as thefollowing equation (16):dV=LIMHL(PMF _(max) −PMF)*(1/(1+sτ)*K   (16),where LIMHL( )denotes a function that limits a value in the parenthesisto the range between the upper deviation-limit set value LIMH and thelower deviation-limit set value LIML, and τ denotes a first-order delaytime constant. The time constant τ is 10 ms to 100 ms orders ofmagnitude.

Preferable settings of the modulation index set value PMF_(max), theupper deviation-limit set value LIMH, and the lower deviation-limit setvalue LIML are as follows for Embodiment. The modulation index set valuePMF_(max) is preferably set to 1.0. This is because that, at the timewhen a modulation index PMF reaches 1.0, i.e., an output voltage of theinverter 2 reaches its maximum voltage, an input to the limiter 81becomes zero or less, so that a negative corrected current command dVcan be generated, which is preferable for performing a field-weakenedcontrol while the output voltage of the inverter 2 is maximized.

The upper deviation-limit set value LIMH is preferably set to a valuethat is obtained by dividing by the gain K the maximum d-axis currentId_(max) (referred to as “maximum field-weakening current”), which iscalculated in advance, required to flow through the electric motor 6when producing a desired torque command T*, taking into account afluctuation range of the capacitor voltage EFC. For example, when themaximum d-axis current Id_(max) is 100 A and the gain K is set to100,000, the upper deviation-limit set value LIMH becomes 0.001. Thelower deviation-limit set value LIML is preferably set to zero. By thussetting the set values, when a modulation index PMF is 1.0 or smaller,i.e., when there is a margin between a voltage command and the maximumoutput voltage of the inverter 2, the corrected current command dV isnot output. At the time when the modulation index PMF exceeds 1.0, i.e.,when a voltage command slightly exceeds the maximum output voltage ofthe inverter 2, the limiter 81 generates a negative output value, sothat a corrected current command dV is output. An unnecessary d-axiscurrent id therefore does not flow, which allows a current through theelectric motor 6 to be minimized.

By thus generating a corrected current command dV based on themodulation index PMF that is a value obtained by normalizing themagnitude of a command output-voltage vector for the inverter by thevoltage EFC of the capacitor 1, an appropriate corrected current commanddV can be obtained, independently of the magnitude of the voltage EFC ofthe capacitor 1, depending on an excess ratio of the commandoutput-voltage vector magnitude for the inverter to the maximum voltagethat the inverter 2 is able to output. Accordingly, stable operation canalso be obtained in application to electric railcars whose voltage EFCof the capacitor 1 fluctuates.

Furthermore, by generating the corrected current command dV using thecombination of the proportional gain element 83 and the first-orderdelay element 82, a stable operation can be performed even when theelectric motor 6 falls into an operation range where the field-weakeningcontrol is not theoretically applied, for example, when a torque commandT* is excessive for rotation speed of the electric motor 6. In such asituation, even though the d-axis current command id* is corrected to anegative by a corrected current command dV, the magnitude of the commandoutput-voltage vector for the inverter cannot be reduced to the maximumvoltage or less that the inverter is able to output. Namely, with thecombination of the proportional gain element 83 and the first-orderdelay element 82, even in the situation where a modulation index PMFremains more than 1.0, a final value of the corrected current command dVnever continue to increase to an excessively large value in theconfiguration of the invention, since the correcting value dV settles toan appropriate value determined from the modulation index PMF, the upperdeviation-limit set value LIMH, and the gain K. In other words, evenwhen the torque command T* is excessive, an appropriate field-weakeningcontrol can be performed.

In a case of a configuration made with a proportional-integralcontroller having an integral element as seen in conventionalconfiguration examples, instead of the above-mentioned combination ofthe gain K and the first-order delay element 82, when a modulation indexPMF remains larger than 1.0, an integration value is accumulated in theintegral element and a corrected current command dV continues toincrease to an excessively large value as time elapses, so that theelectric motor 6 cannot be properly controlled. Moreover, even when themotor recovers from such an uncontrolled state to the normal state, ittakes time to decrease the excessively accumulated integration value toa proper value, which brings poor control during this interim period.For that reason, a complicated operation is required in a practice use,such as setting of an upper limit for the integration value or resettingof the integration value at a predetermined timing.

According to the invention, on the contrary, no such complicatedoperation is needed to perform a stable field-weakening control.

Next, a description is made of a configuration of the parameter-errorcorrection unit 85 that is a key component to demonstrate the effects ofthe invention. The parameter-error correction unit 85 calculates thecontrol phase-angle correcting value dTHV from the d-axis current id,the q-axis current iq, the d-axis current command id*, and the q-axiscurrent command iq*, based on the following equation (17):dTHV=(K ₅ +K ₆ /s)*((id* ² +iq* ²)−(id ² +iq ²))  (17),where K₅ and K₆ denote a proportional gain and an integral gain,respectively, whereby the correction unit operates as aproportional-integral controller.

The first term on the right-hand side of the equation (17) expresses thesum of squares of the d-axis current command id* and the q-axis currentcommand iq*, and denotes the square of the magnitude of the currentcommand vector. The second term on the right-hand side expresses the sumof squares of the d-axis current id and the q-axis current iq, anddenotes the square of the magnitude of the current vector.

The d-axis current id and the q-axis current iq that are electric motorcurrents may sometimes deviate from the d-axis current command id* andthe q-axis current command iq* that are current commands, respectively,by permanent-magnet flux φ_(a) variation and electric-motor parametervariation due to temperature rise of and current through the electricmotor. In this case, by subtracting the square of the current vectormagnitude from that of the current command vector magnitude and bytaking a proportional-integral control of the subtraction result, thecontrol phase angle θ is corrected using the control phase-anglecorrecting value dTHV corresponding to the deviation. An operation canthereby be performed so that the electric motor current comes intocoincidence with the current command, which allows preventing the torqueT of the electric motor 6 from deviating from the torque command T*therefor.

Instead of using the equation (17), the control phase-angle correctingvalue dTHV may be calculated using an equation (18):dTHV=(K ₅ +K ₆ /s)*(SQRT(id* ² +iq* ²)−SQRT(id ² +iq ²))  (18).

The first term on the right-hand side of the equation (18) expresses thesquare root of the sum of squares of the d-axis current command id* andthe q-axis current command iq*, and denotes the magnitude of the currentcommand vector. The second term on the right-hand side expresses thesquare root of the sum of squares of the d-axis current id and theq-axis current iq, and denotes the magnitude of the current vector.

The d-axis current id and the q-axis current iq that are electric motorcurrents may sometimes deviate from the d-axis current command id* andthe q-axis current command iq* that are current commands, respectively,by permanent-magnet flux φ_(a) variation and electric-motor parametervariation due to temperature rise of and current through the electricmotor. In this case, by subtracting the current vector magnitude fromthe current command vector magnitude and by taking aproportional-integral control of the subtraction result, the controlphase angle θ is corrected using the control phase-angle correctingvalue dTHV corresponding to the deviation. An operation can thereby beperformed so that the electric motor current comes into coincidence withthe current command, which allows preventing the torque T of theelectric motor 6 from deviating from the torque command T* therefor.

In addition, since the equation (18) is a complicated equation owing toincluding two square-root operations in comparison with the equation(17), the calculation takes time and involves a significant load on amicrocomputer. Accordingly, it is preferable to use the equation (17).

The difference between a control phase-angle correcting value dTHVcalculated using the equation (17) and that calculated using theequation (18) is explained below. FIG. 6 is a graph showing a relation(obtained using the equation (17)) of the deviation between the sum ofsquares of dq-axis current commands, and that of squares of dq-axiscurrents to the d-axis current error, in Embodiment 1 of the invention.FIG. 7 is a graph showing a relation (obtained using the equation (18))of the deviation between the magnitude of the current command vector andthat of the current vector to the d-axis current error, in Embodiment 1of the invention. FIG. 8 is a graph showing a relation (obtained usingthe equation (17)) of the deviation between the sum of squares ofdq-axis current commands, and that of squares of dq-axis currents to theq-axis current error, in Embodiment 1 of the invention. FIG. 9 is agraph showing a relation (obtained using the equation (18)) of thedeviation between the magnitude of the current command vector and thatof the current vector to the q-axis current error, in Embodiment 1 ofthe invention.

In FIGS. 6 and 7, respectively shown are a relation of the deviation(vertical axis) of the sum of squares of dq-axis currents from that ofsquares of dq-axis current commands, and a relation of the deviation(vertical axis) of the current vector magnitude from the current commandvector magnitude, to a d-axis current error Δid (horizontal axis), whenthere is an error between the d-axis current id and the d-axis currentcommand id* in a situation of the q-axis current iq being equal to theq-axis current command iq*, i.e., in a situation of the q-axis currenterror being zero. Here, the d-axis current error Δid denotes thesubtraction of the d-axis current id from the d-axis current commandid*.

As shown in FIGS. 6 and 7, it is found that both deviations have asimilar characteristic such that they are substantially linear to thed-axis current error Δid in a range thereof being small (within ±50 A)although the vertical scales are different. In addition, the differenceof the vertical scales is insignificant since it can he adjusted by thegain K5 in the equation (17).

In FIGS. 8 and 9, respectively shown are a relation of the deviation(vertical axis) of the sum of squares of dq-axis currents from that ofsquares of dq-axis current commands, and a relation of the deviation(vertical axis) of the current vector magnitude from the current commandvector magnitude, to a q-axis current error Δiq (horizontal axis), whenthere is an error between the q-axis current iq and the q-axis currentcommand iq* in a situation of the d-axis current id being equal to thed-axis current command id*, i.e., in a situation of the d-axis currenterror being zero. Here, the q-axis current error Δiq denotes thesubtraction of the q-axis current command iq* from the d-axis currentiq.

As shown in FIGS. 8 and 9, it is found that both deviations have asimilar characteristic such that the deviations are substantially linearto the q-axis current error Δiq in a range thereof being small (within±50 A) although the vertical scales are different. In addition, thedifference of the vertical scales is insignificant since it can beadjusted by the gain K5 in the equation (17).

As described above, by using the equation (17), the control phase-anglecorrecting value dTHV can be calculated without lengthening thecalculation time nor involving a significant load on the microcomputer.

FIGS. 10 and 11 illustrate charts showing simulated operating waveformsin Embodiment 1 of the invention. In FIG. 10, simulated operatingwaveforms of torque commands, torques, d-axis current commands, d-axiscurrents, q-axis current commands, and q-axis currents are illustrated,and in FIG. 11, simulated operating waveforms of modulation indexs,corrected current commands, U-phase voltage commands, synchronousthree-pulse PWM mode flags, synchronous one-pulse mode flags, andU-phase currents. As illustrated in FIGS. 10 and 11, it is found that astable operation is achieved during a power operation (during a timeinterval of 0 sec to 2.5 sec) and a regenerative operation (during atime interval of 2.7 sec to 5.3 sec), which is described in detailbelow.

During the time from 0 sec to around 0.7 sec, a voltage applied to theelectric motor 6 and also the modulation index PMF linearly increase,and the multi-pulse PWM mode (its mode flag is not indicated in thefigures) and control mode 1 are selected.

Since the modulation index PMF equals to or exceeds a predeterminedvalue at the time around 0.7 sec, the synchronous three-pulse PWM modeand control mode 2 are selected. During the time from 0.7 sec to around1.0 sec, the modulation index PMF further linearly increases but itsmagnitude is smaller than 1.0.

In addition, the magnitude of the U-phase voltage command Vu* decreasesimmediately after the change to the synchronous three-pulse PWM mode atthe time around 0.7 sec, this is due to the command voltage magnitudePMFM that has been multiplied by 1.274 by the gain adjustment table 54in the multi-pulse PWM mode is change to be multiplied by 1.0 asdescribed above.

During from the boot-up to the time around 1.0 sec, the maximum torquecontrol is taken by the current command generation unit 10, and thed-axis current command id* and the q-axis current command iq* areconstant because the torque command T* is constant.

At the time around 1.0 sec, since the modulation index PMF reaches 1.0,the synchronous one-pulse mode is selected as the pulse mode and thecorrected current command dV negatively increases, so that the d-axiscurrent command id* further increases negatively accordingly. The d-axiscurrent id follows the d-axis current command id*, to negativelyincrease. From that, it is found that the field-weakening control isdesirably performed as well as the modulation index PMF is kept at avalue infinitely close to 1.0, that is, the terminal voltage of theelectric motor 6 is kept constant.

From the fact that the torque follows the torque command T*, it is foundthat the electric motor 6 stably accelerates its rotation speed becausethe torque command T* is reduced in inverse proportion to the rotationspeed in order that the electric motor 6 operates to output constantpower.

At the time around 1.8 sec, the torque command T* is once reduced tozero to stop the inverter 2 (the gate signals U, V, W, and X, Y, Z areall switched off). Then, the inverter is rebooted in a power operationmode at the time around 2.0 sec and operated in the power operation modetill the time around 2.5 sec. It is found that the torque T is also incoincidence with the torque command T* during the series of suchoperations, proving that the normal operation is performed.

Moreover, since the pulse modes are switched over depending on themodulation index PMF, it is found, from the flags of the synchronousthree-pulse PWM mode and those of the synchronous one-pulse PWM mode,that the pulse mode is automatically changed to the synchronousthree-pulse PWM mode when the modulation index PMF becomes smaller than1.0 during the processes of the reduction of the torque command T* andof the reboot.

During the time from around 2.2 sec to 2.3 sec, the torque command T*becomes large with respect to the rotation speed, so that the electricmotor 6 is operated in a range where the field-weakening control thereofis not theoretically realized. In that range, even though the d-axiscurrent command id* is corrected to a negative using the correctedcurrent command dV, the magnitude of the command output-voltage vectorfor the inverter cannot be reduced to the maximum voltage or less thatthe inverter is able to output. Since the corrected current command dVis however limited to those within a constant value (−150 A) that isdetermined from the modulation index PMF, the upper deviation-limit setvalue LIMH and the gain K, it is found that the corrected currentcommand dV does not become excessively large.

At the time around 2.7 sec, the torque command T* is set to be negative,and the inverter is booted in the regenerative operation mode. At thetime around 3.2 sec, the torque command T* is once set to zero to stopthe inverter 2 (the gate signals U, V, W, and X, Y, Z are all switchedoff) and then the inverter is rebooted at the time around 3.4 sec. It isfound that the torque T is also in coincidence with the torque commandT* during the series of such operations, proving that the normaloperation is performed.

It is also found that since the modulation index PMF becomes smallerthan 1.0 during the processes of increasing and reducing the torquecommand T*, the pulse mode is automatically changed to the synchronousthree-pulse PWM mode, and that the synchronous one-pulse PWM mode isautomatically selected at the stage when the modulation index PMFreaches 1.0.

While the regenerative operation is continuously performed after thetime around 3.4 sec, the d-axis current command id* is adjusted to benegative using the corrected current command dV, so that thefield-weakening control is normally performed till the time around 4.2sec.

After the time around 4.2 sec, since the terminal voltage of theelectric motor 6 decreases owing to rotation speed reduction thereof,the modulation index PMF becomes less than 1.0 and the corrected currentcommand dV automatically becomes zero. At the same time, the pulse modeis changed to the synchronous three-pulse PWM mode. With furtherdecrease of the modulation index PMF at the time around 4.5 sec, thepulse mode is changed to the multi-pulse PWM mode and the control mode 1is simultaneously selected.

In this way, it is found that the stable operations can be performedeven in the field-weakening operation range and the transitions betweenthe field-weakening operation range and the other ranges are also stablyachieved. It is further found that the transitions between the controlmodes and between the pulse modes can be stably achieved.

As described above, the present invention can provide a vectorcontroller for a permanent-magnet synchronous electric motor that canperform stable transitions in operation mode under a range from a low toa high rotation speed of the electric motor 6 with pulse modes andcontrol modes of the inverter 2 being switched over, and can perform,with a configuration more simplified than conventional ones, a stablefield-weakening operation in a one-pulse mode in which output voltage ofthe inverter 2 can be maximized in the high speed range.

The pulse modes and the control modes may be switched over based not ona modulation index but on a voltage command, motor frequency, inverterfrequency, railcar speed, or the like.

The configuration described in Embodiment is an exemplar of the subjectmatter of the present invention and can be combined with another priorart. Modifying the configuration, for example, omitting part thereof canalso be made within the scope of the invention.

While the subject matter of the invention has been described using anapplication to a controller for electric railcars in the specification,applicable fields are not limited to this. The invention can be appliedto various related fields such as electric vehicles and elevators.

1. A vector controller for a permanent-magnet synchronous electric motorthat controls an alternating current from an inverter that drives thepermanent-magnet synchronous electric motor so as to come intocoincidence with a current command, comprising: a reference phase-anglecalculation unit for generating a reference phase angle of thepermanent-magnet synchronous electric motor; a current commandgeneration unit for generating the current command using a given torquecommand; a current control unit for making a control calculation of acurrent error between the current command and a current through thepermanent-magnet synchronous electric motor, to output the calculatedcurrent error; a decoupling voltage calculation unit for calculating afeed-forward voltage using motor parameters of the permanent-magnetsynchronous electric motor and the current command; a modulation indexcalculation unit for outputting a modulation index for the inverter byreceiving a direct-current voltage to the inverter and a voltage commandthat is the sum of the current error and the feed-forward voltage; acontrol phase-angle calculation unit for outputting a control phaseangle for the inverter by receiving the voltage command and thereference phase angle; a pulse-width-modulation signal generation unitfor generating pulse-width-modulation signals for the inverter using themodulation index and the control phase angle; and a current commandcompensator for correcting the current command using a corrected currentcommand calculated based on the modulation index; wherein the currentcommand compensator sets the corrected current command to a valueobtained by processing through a time delay element and by multiplyingby a predetermined constant the difference between the modulation indexand a predetermined modulation index set value.
 2. The vector controllerfor a permanent-magnet synchronous electric motor of claim 1, whereinthe current command compensator, before multiplying by the predeterminedconstant the difference between the modulation index and thepredetermined modulation index set value, limits the difference to arange between an upper and a lower deviation-limit set values.
 3. Thevector controller for a permanent-magnet synchronous electric motor ofclaim 2, wherein the upper deviation-limit set value is larger than zeroand the lower-deviation-limit set value is equal to or smaller thanzero.
 4. The vector controller for a permanent-magnet synchronouselectric motor of claim 2, wherein the upper deviation-limit set valueis set based on a maximum field-weakening current necessary for thepermanent-magnet synchronous electric motor to generate the commandtorque within a range of variation in the direct-current voltage to theinverter.
 5. The vector controller for a permanent-magnet synchronouselectric motor of claim 1, wherein the modulation index is defined ashaving a value of unity when the inverter outputs square waves whosefundamental wave component of line voltage of the inverter reaches amaximum.
 6. The vector controller for a permanent-magnet synchronouselectric motor of claim 1, wherein the modulation index set value is setto a modulation index by which the inverter outputs square waves whosefundamental wave component of line voltage of the inverter reaches amaximum.
 7. The vector controller for a permanent-magnet synchronouselectric motor of claim 1, wherein the pulse-width-modulation signalgeneration unit switches over pulse modes of the inverter depending onthe modulation index.
 8. The vector controller for a permanent-magnetsynchronous electric motor of claim 1, wherein thepulse-width-modulation signal generation unit can hold a carrier signalat zero depending on the modulation index.
 9. The vector controller fora permanent-magnet synchronous electric motor of claim 1, furthercomprising a parameter-error correction unit for calculating, from thecurrent command and the current through the permanent-magnet synchronouselectric motor, a control phase-angle correcting value used forcorrecting the control phase angle.
 10. The vector controller for apermanent-magnet synchronous electric motor of claim 9, wherein thecontrol phase-angle correcting value is calculated, by computing thevector control in a rotating coordinate system having a d-axis and aq-axis orthogonal to each other, based on the sum of squares of a d-axisand a q-axis components of the current command, and on that of squaresof a d-axis and a q-axis components of the current through thepermanent-magnet synchronous electric motor.
 11. The vector controllerfor a permanent-magnet synchronous electric motor of claim 10, whereinthe control phase-angle correcting value is calculated by taking aproportional-integral control of the difference between the sum ofsquares of a d-axis and a q-axis components of the current command, andthe sum of squares of a d-axis and a q-axis components of the currentthrough the permanent-magnet synchronous electric motor.
 12. The vectorcontroller for a permanent-magnet synchronous electric motor of claim 9,wherein the parameter-error correction unit determines whether to makethe calculation based on a predetermined signal.
 13. The vectorcontroller for a permanent-magnet synchronous electric motor of claim12, wherein the predetermined signal is the modulation index.
 14. Thevector controller for a permanent-magnet synchronous electric motor ofclaim 13, wherein the pulse-width-modulation signal generation unitswitches over pulse modes of the inverter depending on the modulationindex, and when the modulation index is higher than that correspondingto a lower limit at which a synchronous three-pulsepulse-width-modulation mode is selected for the inverter, the currentcontrol unit does not make its calculation and the parameter-errorcorrection unit makes its calculation instead, and when the modulationindex is lower than that corresponding to the lower limit at which thesynchronous three-pulse pulse-width-modulation mode is selected for theinverter, the current control unit makes its calculation and theparameter-error correction unit does not make its calculation.
 15. Thevector controller for a permanent-magnet synchronous electric motor ofclaim 1, wherein the current control unit determines whether to make thecalculation based on a predetermined signal.
 16. The vector controllerfor a permanent-magnet synchronous electric motor of claim 15, whereinthe predetermined signal is the modulation index.